uTracer NXT — Vol 3: Measurement Theory

Why pulse instead of holding HV continuously — and how the NXT charges, switches, and samples

Figure 1 — The three-phase pulsed cycle: charge the reservoir cap slowly, fire the HV switch for ~1 ms while the grid is biased, sample the current, then discharge. Source: hand-authored SVG.

3.1 The problem pulsing solves

Put a real operating point on a power tube — say 300 V at 100 mA — and the plate is dissipating 30 W. Hold that continuously across a full sweep of operating points and you need a bench supply that can source that power, a heatsink to get rid of it, and you risk cooking the tube if you wander past its rated plate dissipation (Pa,max). That is the bulky, expensive, dangerous instrument the uTracer was invented to avoid.

The trick is that a tube doesn’t care whether you hold the operating point for a second or a millisecond — the current at a given (Va, Vg) is the same. So the uTracer establishes the operating point for only about 1 ms, samples the current at the end of that window, and disconnects. The average power is the peak power scaled by the duty cycle, and with a ~1 ms pulse fired only when you ask for a point, that duty cycle is minute. 30 W of instantaneous plate dissipation becomes milliwatts of average heating. This is the single idea behind every uTracer, the NXT included.

3.1.1 Why that lets you safely exceed Pa,max

Plate-dissipation ratings are thermal limits — they describe how much heat the anode can shed continuously before it overheats and outgasses or melts. A tube’s thermal mass means the anode temperature responds to average power over many milliseconds, not to a single 1 ms spike. Because the pulsed operating point deposits only a tiny slug of energy (power × 1 ms) and then the tube sits cold until the next point, the anode never heats up the way continuous operation would. That is why a uTracer can legitimately trace curves at instantaneous power well above the tube’s continuous Pa,max — the uTracer6 demonstrated this dramatically, pulsing an EL34 to roughly 900 V and 900 mA (near 1 kW instantaneous) with the tube surviving. The NXT operates in a lower envelope but on the identical thermal principle.

3.2 The three phases of a measurement

The timing diagram above shows one full cycle. It has three phases, and it is worth being precise about the timescales because they differ by orders of magnitude.

Phase 1 — Charge (seconds). The plate voltage is not switched from a big supply; it is stored first. The boost converter (Vol 2) pumps the 330 µH inductor in short bursts and dribbles charge into a 100 µF / 500 V reservoir capacitor until it reaches the target plate voltage. Because the boost is small and the cap is large, this takes on the order of a few seconds to reach ~500 V. During this whole time the tube sees nothing.

Phase 2 — Pulse and sample (~1 ms). When the cap is charged, the PIC closes the high-voltage switch (the NMOS switch inherited from the uTracer6) to connect the charged cap to the tube’s plate. Simultaneously the grid bias amplifier is enabled so the correct control-grid voltage is present only during the pulse. Plate and screen current now flow, pulling the reservoir cap down slightly, and return through the sense resistor. At the end of the ~1 ms window the ADC samples the sense-resistor voltages — that captured value is the anode current Ia and screen current Is for this (Va, Vs, Vg) point.

Phase 3 — Discharge / idle. The HV switch opens, the grid amplifier is disabled, and residual charge on the switch’s parasitic capacitance and the rail is bled off so the next point starts clean. Then the cycle repeats for the next setpoint.

Figure 2 — Two switches, two jobs: the boost NMOS charges the reservoir cap over seconds; the HV switch NMOS connects it to the tube for the ~1 ms pulse, and the plate current returns through the 14.3 Ω shunt… — Figure 2 — Two switches, two jobs: the boost NMOS charges the reservoir cap over seconds; the HV switch NMOS connects it to the tube for the ~1 ms pulse, and the plate current returns through the 14.3 Ω shunt. Source: hand-authored SVG.

3.3 How the high voltage is generated: the two-switch scheme

It is easy to conflate the two NMOS transistors in the design, so the second figure separates them. There are genuinely two switches doing two different jobs:

The boost switch is the one inside the switched converter. Driven by the PIC’s PWM, it repeatedly grounds one end of the 330 µH inductor; when it opens, the inductor’s collapsing field forces current through a diode into the reservoir cap, ratcheting the cap voltage up. Do this thousands of times and the cap climbs to hundreds of volts from a 20 V input. This is a boost (step-up) converter, not a switched-capacitor charge pump — the energy store is the inductor’s magnetic field, and the reservoir cap is what holds the result.
The HV switch is the one that connects that already-charged cap to the tube for the measurement pulse. This is the device that has to hold off the full plate voltage when open and pass the plate current cleanly when closed, which is exactly why moving it from a fragile PNP to a robust NMOS (the uTracer6 advance) mattered so much.

The NXT uses a 700 V-class NMOS for the high-voltage switch, appropriate to its ~500 V envelope and driven from a low-voltage logic-level gate drive, where the kilovolt uTracer6 needed 1000 V+ devices (and, for its boost switch, an exotic 1700 V silicon-carbide MOSFET). Lower target voltage is what lets the NXT use cheaper, more available parts than the 6 while keeping the same architecture.

3.4 How Ip and Is are actually sampled

The current measurement is a chain, and each stage earns its place:

Shunt. Anode and screen currents each flow through a sense resistor (14.3 Ω default). Ohm’s law turns current into a small voltage — 100 mA across 14.3 Ω is 1.43 V, a comfortable signal; 1 mA is 14.3 mV, a small one.
Zero-drift buffer. That shunt voltage is buffered by the MCP6V86 zero-drift op-amp. Its tiny offset (tens of microvolts) is what makes the small currents trustworthy — a few millivolts of amplifier offset would swamp a milliamp-scale reading, which is why the OPA227’s low offset mattered and why its modern replacement had to match it.
Programmable gain. The PGA113 then multiplies by a software-chosen factor (1× to 200×) so the signal fills the ADC’s input range regardless of whether the tube is drawing microamps or hundreds of milliamps. This is effectively an autoranging front end: the software picks the gain, records which gain it used, and scales the result.
ADC. Finally the PIC’s 10-bit ADC digitizes the amplified voltage. With the PGA supplying the range and the zero-drift buffer supplying the floor, 10 bits per range covers a wide dynamic span. The firmware can also average multiple pulses per point to beat down noise.

The result of one pulse, then, is a pair of digitized currents plus the actually-achieved voltages, which the host converts back to engineering units using the calibration constants (Vol 4).

The NXT doesn’t reinvent the pulsed principle — it refines its execution:

The NMOS HV switch from the uTracer6, replacing the obsolescing/fragile PNP switch, so the everyday tracer inherits the kilovolt model’s fault-robustness (Vol 2).
A PGA-based front end (MCP6V86 + PGA113) in place of the fixed OPA227 sense amplifier, giving clean software-controlled ranging built from current-production parts.
A simplified rail scheme (+5 V and −105 V, no ±15 V auxiliaries) and a pulsed, optocoupler-gated grid amplifier that is powered only during the measurement window — lower dissipation and inherent fault protection.

None of this changes the headline theory: charge slowly, pulse briefly, sample at the end, and let duty cycle keep the tube cool. The comparison in Vol 5 puts the three generations side by side; Vol 4 gets the kit built and calibrated so the numbers coming out of this chain are trustworthy.